Magnetic Field Sensors and Associated Methods for Removing Undesirable Spectral Components

ABSTRACT

Magnetic field sensors and associated techniques use a Hall effect element in a chopping arrangement in combination with a feedback path configured to reduce undesirable spectral components generated by the chopping.

CROSS REFERENCE TO RELATED APPLICATIONS

Not Applicable.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

Not Applicable.

FIELD OF THE INVENTION

This invention relates generally to magnetic field sensors and, moreparticularly, to a magnetic field sensor having a Hall element andelectronics to reduce undesirable spectral components generated when theHall Effect element is chopped.

BACKGROUND OF THE INVENTION

As is known, there are a variety of types of magnetic field sensingelements, including, but not limited to, Hall Effect elements,magnetoresistance elements, and magnetotransistors. As is also known,there are different types of Hall Effect elements, for example, planarHall elements, vertical Hall elements, and circular vertical Hallelements (CVH). As is also known, there are different types ofmagnetoresistance elements, for example, anisotropic magnetoresistance(AMR) elements, giant magnetoresistance (GMR) elements, tunnelingmagnetoresistance (TMR) elements, Indium antimonide (InSb) elements, andmagnetic tunnel junction (MTJ) elements.

Hall Effect elements generate an output voltage proportional to amagnetic field. In contrast, magnetoresistance elements changeresistance in proportion to a magnetic field. In a circuit, anelectrical current can be directed through the magnetoresistanceelement, thereby generating a voltage output signal proportional to themagnetic field.

Magnetic field sensors, which use magnetic field sensing elements, areused in a variety of applications, including, but not limited to, acurrent sensor that senses a magnetic field generated by a currentcarried by a current-carrying conductor, a magnetic switch (alsoreferred to herein as a proximity detector) that senses the proximity ofa ferromagnetic or magnetic object, a rotation detector that sensespassing ferromagnetic articles, for example, gear teeth, and a magneticfield sensor that senses a magnetic field density of a magnetic field.Particular magnetic field sensor arrangements are used as examplesherein. However, the circuits and techniques described herein apply alsoto any magnetic field sensor.

It is known that Hall Effect elements exhibit an undesirable DC offsetvoltage. Techniques have been developed to reduce the DC offset voltage,while still allowing the Hall Effect element to sense a DC magneticfield. One such technique is commonly referred to as “chopping.”Chopping is a technique by which a Hall Effect element is driven in twoor more different directions, and outputs are received at differentoutput terminals as the Hall Effect element is differently driven. Withchopping, offset voltages of the different driving arrangements tend tocancel toward zero.

However, the chopping tends to generate undesirable spectral components(i.e., frequency components in the frequency domain). The undesirablespectral components can be removed with filters.

Circuits that both chop a Hall element and that use one or more filtersto remove undesirable spectral components is described in U.S. patentapplication Ser. No. 13/095,371, filed on Apr. 27, 2011, entitled“Circuits and Methods for Self-Calibrating or Self-Testing a MagneticField Sensor,” assigned to the assignee of the present invention, andwhich is incorporated by reference herein in its entirety.

While convention arrangements that use filters can effectively reducethe undesirable spectral components, it will be understood that thefilters tend to reduce a bandwidth or a response time of the magneticfield sensors that use filters.

It would be desirable to provide a magnetic field sensor that uses aHall Effect element in a chopping arrangement, and that can reduceundesirable spectral components generated by the chopping, but that doesnot reduce a bandwidth or response time of the magnetic field sensor.

SUMMARY OF THE INVENTION

The present invention provides a magnetic field sensor that uses a HallEffect element in a chopping arrangement, and that can reduceundesirable spectral components generated by the chopping, but that doesnot reduce a bandwidth or response time of the magnetic field sensor.

In accordance with one aspect of the present invention, a magnetic fieldsensor includes a magnetic field sensing element configured to generatean electronic signal in response to a magnetic field. The magnetic fieldsensor also includes an N-phase modulator coupled to receive theelectronic signal and configured to generate a modulated signal having aplurality of frequency components at different frequencies. At least twoof the plurality of frequency components correspond to at least twoundesirable frequency components and one of the plurality of frequencycomponents corresponds to a desirable frequency component. The desirablefrequency component comprises a magnetic field signal representative ofthe magnetic field. The magnetic field sensor also includes a primarycircuit path coupled to receive the modulated signal and to process themodulated signal to generate an output signal representative of themagnetic field signal. The magnetic field sensor also includes afeedback circuit path coupled at both ends of the feedback circuit pathto the primary circuit path and forming a feedback loop. The feedbackcircuit path is configured to generate a feedback signal and is coupledto add the feedback signal to the primary circuit path to cancel the atleast two undesirable frequency components.

In accordance with another aspect of the present invention, a method ofprocessing a signal in a magnetic field sensor includes generating amagnetic field signal with a magnetic field sensor in response to amagnetic field. The method also includes modulating the magnetic fieldsignal with an N-phase modulator coupled to receive the magnetic fieldsignal and configured to generate a modulated signal having a pluralityof frequency components at different frequencies. At least two of theplurality of frequency components correspond to at least two undesirablefrequency components and one of the plurality of frequency componentscorresponds to a desirable frequency component. The method also includesprocessing the modulating signal with a primary circuit path coupled toreceive the modulated signal. The primary circuit path is configured togenerate an output signal representative of the desirable frequencycomponent. The method also includes generating a feedback signal with afeedback circuit path and adding the feedback signal to the primarycircuit path to cancel the at least two undesirable frequencycomponents.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features of the invention, as well as the invention itselfmay be more fully understood from the following detailed description ofthe drawings, in which:

FIG. 1 is a block diagram showing a Hall Effect element and a switchingcircuit coupled in a four-phase chopping arrangement;

FIG. 2 is a set of graphs showing signals within the circuits of FIG. 1;

FIG. 3 is a block diagram showing an exemplary magnetic field sensorhaving a Hall effect element used in a chopping arrangement, having aprimary signal path, and having a feedback circuit path configured toreduce undesirable spectral components in the primary signal path thatresults from the chopping;

FIG. 4 is a block diagram showing another exemplary magnetic fieldsensor, having a primary signal path, and having a Hall effect elementused in a chopping arrangement and having a feedback circuit pathconfigured to reduce undesirable spectral components that results fromthe chopping; and

FIG. 5 is a block diagram of a four-phase switched capacitor notchfilter integrator circuit;

FIG. 5A is a series of graphs showing clock signals associated with thefour-phase switched capacitor notch filter integrator circuit of FIG. 5;

FIG. 5B is a block diagram of a conventional switched capacitorintegrator circuit;

FIG. 5C is a series of graphs showing clock signals associated with theswitched capacitor integrator circuit of FIG. 5B;

FIG. 5D is a graph showing an exemplary transfer function that can beassociated with the four-phase switched capacitor notch filterintegrator circuit of FIG. 5;

FIGS. 6-23 graphs showing spectral content of signals at various pointswithin the magnetic field sensors of FIGS. 3 and 4, in particular, for afour-phase chopping arrangement and for the magnetic field sensed by themagnetic field sensors that contains only a DC component;

FIG. 24 is a graph showing a time behavior of an output signal generatedby the magnetic field sensors of FIGS. 3 and 4 for a time periodbeginning at startup of the magnetic field sensors and for the magneticfield sensed by the magnetic field sensors the contains only the DCcomponent;

FIG. 25 is a graph showing a time behavior of one of the signals withinthe magnetic field sensors of FIGS. 3 and 4 for a time period beginningat startup; and

FIG. 26 is a graph showing a time behavior of another one of the signalswithin the magnetic field sensors of FIGS. 3 and 4 for a time periodbeginning at startup.

DETAILED DESCRIPTION OF THE INVENTION

Before describing the present invention, some introductory concepts andterminology are explained. As used herein, the term “magnetic fieldsensing element” is used to describe a variety of types of electronicelements that can sense a magnetic field. The magnetic field sensingelements can be, but are not limited to, Hall Effect elements,magnetoresistance elements, or magnetotransistors. As is known, thereare different types of Hall Effect elements, for example, planar Hallelements, vertical Hall elements, and circular vertical Hall (CVH)elements. As is also known, there are different types ofmagnetoresistance elements, for example, anisotropic magnetoresistance(AMR) elements, giant magnetoresistance (GMR) elements, tunnelingmagnetoresistance (TMR) elements, Indium antimonide (InSb) elements, andmagnetic tunnel junction (MTJ) elements.

As is known, some of the above-described magnetic field sensing elementstends to have an axis of maximum sensitivity parallel to a substratethat supports the magnetic field sensing element, and others of theabove-described magnetic field sensing elements tend to have an axis ofmaximum sensitivity perpendicular to a substrate that supports themagnetic field sensing element. In particular, most, but not all, typesof magnetoresistance elements tend to have axes of maximum sensitivityparallel to the substrate and most, but not all, types of Hall elementstend to have axes of sensitivity perpendicular to a substrate.

As used herein, the term “magnetic field sensor” is used to describe acircuit that includes a magnetic field sensing element. Magnetic fieldsensors are used in a variety of applications, including, but notlimited to, a current sensor that senses a magnetic field generated by acurrent carried by a current-carrying conductor, a magnetic switch (alsoreferred to herein as a proximity detector) that senses the proximity ofa ferromagnetic or magnetic object, a rotation detector that sensespassing ferromagnetic articles, for example, gear teeth, and a magneticfield sensor (e.g., a linear magnetic field sensor) that senses amagnetic field density of a magnetic field. Linear magnetic fieldsensors are used as examples herein. However, the circuits andtechniques described herein apply also to any magnetic field sensorcapable of detecting a magnetic field.

As used herein, the term “magnetic field signal” is used to describe anycircuit signal that results from a magnetic field experienced by amagnetic field sensing element.

While circuits are described below that use Hall elements in choppedarrangements and that have certain circuits to reduce undesirablespectral components resulting from the chopping, similar techniques canbe used with other magnetic field sensors that support N-phase chopping,in order to reduce undesirable spectral components no matter what theirsource.

Referring to FIG. 1, an exemplary Hall Effect element 10 is chopped byoperation of an N-Phase modulator circuit 12, here shown to be a fourphase modulator circuit 12. The four phase modulator circuit 12 caninclude a plurality of switches coupled to the Hall Effect element 10 infour different arrangements. The plurality of switches can provide adifferential output signal 14 a, 14 b.

In operation, and in accordance with four-phase (i.e., 4×) chopping, theHall Effect element 10 can be driven with currents in four differentdirections represented by four arrows numbered 1-4. At the same time,respective pairs of the switches are closed. For example, when thecurrent through the Hall Effect element 10 is in a direction representedby the arrow labeled 1, two switches also labeled 1 are closed and allof the other switches are open. There are four such arrangements ofdrive currents and pairs of switch closures indicated. For sucharrangements occur in sequence, and thus, they are often referred to aschopping phases. The chopping phases are sequenced at a rate related toa chopping frequency, described more fully below.

It will be understood that each chopping phase in sequence contributesto the differential output signal 14 a, 14 b. It will also be understoodthat by proper sequencing of the drive currents and the switch pairclosures, some components of the resulting differential output signal 14a, 14 b can shift to one or more frequencies related to the choppingfrequency. Depending upon phasing, the components that are shifted infrequency include either the DC offset of the Hall Effect element 10 ora signal (Bext) corresponding to a magnetic field sensed by the HallEffect element 10.

In examples described more fully below, a chopping phase sequence isdescribed that results in the signal corresponding to the magnetic fieldsensed by the Hall Effect element 10 being shifted to a frequencyrelated to the chopping frequency.

Referring now to FIG. 2, three signals, 20, 24, 28 are shown in the timedomain. Horizontal axes have scales in units of time in arbitrary unitsand a vertical axis has a scale in units of volts in arbitrary units.

The signal 20 represents a DC offset voltage component of the HallEffect element 10 of FIG. 1, which remains at baseband within thedifferential output signal 14 a, 14 b of FIG. 1. The DC offset voltage20 has a magnitude 22.

The signal 24 corresponds to one of the higher frequency spectralcomponents (occurring at a frequency equal to half of the choppingfrequency, fchop/2 (and also at odd harmonics thereof, at loweramplitude)), but shown as an AC signal in the time domain, of thedifferential output signal 14 a, 14 b generated by the chopping ofFIG. 1. An arrow 26 is indicative of twice the amplitude of thiscomponent.

In operation of the chopping arrangement of FIG. 1, the signals 20 and24 exist together in combination with other spectral components.

The signal 28 includes the DC signal component 20, the AC signalcomponent 24, and also another AC signal component corresponding to themagnetic field sensed by the Hall Effect element 10 of FIG. 1, whichoccurs at a frequency of fchop. An arrow 30 is indicative of twice anamplitude of the AC signal component corresponding to the magnetic fieldsensed by the Hall Effect element 10.

It should be appreciated that the steps of the signal 28 correspond tosamples of signals generated by the Hall element 10 of FIG. 1 within thedifferential output signal 14 a, 14 b of FIG. 1 as the plurality pairsof switches close and open in sequence. As indicated, one period of thesampled signal 28 has a “chopping period,” T_(ch)=1/f_(ch), where f_(ch)is a so-called “chopping frequency,” also referred to herein as fchop.It should be appreciated that the chopping period, T_(ch), correspondsto only two samples out of a four sequential samples provided by theswitching circuit 12 of FIG. 1. Nevertheless, these are the conventionalmeanings of the chopping period and chopping frequency, fchop.

It should be understood that the AC signal component corresponding tothe magnetic field sensed by the Hall effect element 10, i.e., at thefrequency of fchop, is the only desired signal component within thesignal 28, and the DC offset signal 20 and the AC signal component 24,at the frequency of fchop/2, are not desired. Thus it is desirable toremove the DC component 20 and the AC signal component 24 from thesignal 28. Techniques described below can remove the undesirable signalcomponents

Within the signal 28, it should be appreciated that the undesirable ACsignal component 24, as shown, generates a signal component withmagnitude 26. Furthermore, it should be appreciated that the undesirableDC signal component 20 generates a DC signal component with a magnitude22, i.e., an undesirable DC offset of the signal 28.

Thus, it should be appreciated then that the signal 28 has all threesignal components (and corresponding spectral components in thefrequency domain), two of which are undesirable. The signal 28 can bethe same as or similar to the differential signal 14 a, 14 b of FIG. 1.

Referring briefly to FIG. 1, for four-phase chopping as shown, there arefour clock signals controlling four respective pairs of switches withinthe switching circuit 12, and there are four respective directions ofdrive currents within the Hall element 10. With the above definition ofchopping frequency, fchop, each clock has a frequency of 2fchop/N, whereN is a number of phases in the chopping. For four-phase chopping, N=4,and the frequency of each one of the clocks is fchop/2. Referring againto FIG. 2, in other words, the period, T_(ch), at the chopping frequencycorresponds to only two of N sequential samples provided by N-phasechopping.

Referring now to FIG. 3, a magnetic field sensor 50 includes a HallEffect element 52 coupled to an N-phase modulator circuit 54 in achopped circuit 96. The Hall Effect element 52 and the N-phase modulatorcircuit can be the same as or similar to the Hall Effect element 10 andthe N-phase modulator 12 of FIG. 1. A generalized N-phase choppingarrangement is shown in FIG. 3 and is also used in further examplesbelow. It should be understood that many different chopping arrangementscan be used, including, but not limited to, a 2×, a 4×, and an 8×chopping arrangement, where N=2, 4, or 8.

The N-phase modulator circuit 54 can be configured to generate adifferential output signal 54 a, 54 b, which can be the same as orsimilar to the differential output signal 14 a, 14 b of FIG. 1. For easyreference in figures below, the differential signal 54 a, 54 b islabeled as a signal A, and other differential signal described below arelabeled with other letters.

A primary circuit path 112, and, in particular an amplifier 56, can becoupled to receive the differential signal 54 a, 54 b and configured togenerate an amplified differential output signal 56 a, 56 b, which islabeled as a signal B. In some embodiments, the amplifier 56 is atransconductance amplifier.

A summing node 58 can be coupled to receive the signal 56 a and anothersumming node 60 can be coupled to receive the signal 56 b. The summingnodes 58, 60 can also be coupled to receive other signals described morefully below.

The summing nodes 58, 60 are configured to provide a differential signal58 a, 60 a, which is labeled as a signal C.

Another amplifier 62 is coupled to receive the differential signal 58 a,60 a and configured to generate an amplified signal 62 a, 62 b, which islabeled as a signal D. In some embodiments, the amplifier 62 is also atransconductance amplifier.

A switching circuit 64 can be coupled to receive the differential signal62 a, 62 b and configured to generate a differential switched signal 64a, 64 b, which is labeled as a signal E.

Another amplifier 66 is coupled to receive the differential switchedoutput signal 64 a, 64 b and configured to generate another differentialamplified signal 66 a, 66 b, which is labeled as a signal F. In someembodiments the amplifier 66 is a transconductance amplifier.

While transconductance amplifiers are described above, in otherembodiments, the various amplifiers can be voltage amplifiers.

The differential amplified signal 66 a, 66 b can correspond to adifferential output signal from the magnetic field sensor 50. It isdesirable that the differential output signal 66 a, 66 b consist only ofsignal components directly related to that magnetic field which the HallEffect element 52 senses, and not include undesirable signal components,for example, a DC offset component or other signal components describedabove in conjunction with FIG. 2.

The primary circuit path 112 can be used in conjunction with a gainfeedback path 114. In general, the gain feedback path 114 is used tocontrol and stabilize a gain of the primary circuit path 112. The gainfeedback path 114 can include a feedback network 68 coupled to receivethe differential output signal 66 a, 66 b and configured to generate adifferential signal 68 a, 68 b. The feedback network 68 can becomprised, for example, of passive circuit elements, for example,resistors.

The gain feedback path 114 can also include a switching circuit 70coupled to receive the differential signal 68 a, 68 b and configured togenerate a differential switched signal 70 a, 70 b, which is labeled asa signal M. The summing circuit 58 can be coupled to receive the signal70 a, and the summing circuit 60 can be coupled to receive the signal 70b, providing a feedback arrangement.

The magnetic field sensor 50 can also include one or more feedbackcircuits. Here shown are a first feedback circuit 116 and an Mthfeedback circuit 118. Taken together, the feedback circuits 116, 118form a so-called “feedback circuit path,” which is coupled at both endsto the primary circuit path 112, so as to form a feedback loop.

There are N/2 such feedback circuits within the feedback circuit path,where N equals the number of phases in the chopping of the Hall Effectelement 52. Thus, M=N/2. For N=4, i.e., for four phase (4×) chopping,there are two such feedback circuits within the feedback circuit path.However, for 2× chopping, there is only one feedback circuit, i.e., thefeedback circuit 116, within the feedback circuit path. As describedabove, there can be any number of phases in the chopping of the HallEffect element 52, and any resulting number of feedback circuits withinthe feedback circuit path.

The first feedback circuit 116 can include first and second capacitors72, 74, respectively, coupled to receive the differential output signal66 a, 66 b. At opposite ends of the two capacitors 72, 74, adifferential signal 72 a, 74 a is generated, which is labeled as asignal G, and which has no DC signal component, since the DC componentis blocked by the two capacitors 72, 74.

The first feedback circuit 116 can also include a switching circuit 76coupled to receive the differential signal 72 a, 74 and configured togenerate a differential switched signal 76 a, 76 b, which is labeled asa signal H. An integrator 78 is coupled to receive the differentialswitched signal 76 a, 76 b and configured to generate a differentialintegrated signal 78 a, 78 b, which is labeled as a signal L. Thesumming circuit 58 can be coupled to receive the signal 78 b and thesumming circuit 60 can be coupled to receive the signal 78 a, or viceversa.

The Mth feedback circuit 118 can include a switching circuit 82 coupledto receive the differential signal 76 a, 76 b and configured to generatea differential switched signal 82 a, 82 b, which is labeled as a signalI. An integrator 84 can be coupled to receive the differential switchedsignal 82 a, 82 b and configured to generate a differential integratedsignal 84 a, 84 b, which is labeled as a signal J. A switching circuit86 can be coupled to receive the differential integrated signal 84 a, 84b and configured to generate a differential switched signal 86 a, 86 b,which is labeled as a signal K. The summing circuit 58 can be coupled toreceive the signal 86 b and the summing circuit 60 can be coupled toreceive the signal 78 a, or vice versa.

It should be recognized that the differential signal 78 a, 78 b and thedifferential switched signal 86 a, 86 b are added to signals within theprimary circuit path 112. It will become apparent from discussion belowthat the differential signal 78 a, 78 b can cancel some undesirablesignal components within the primary circuit path 112 and, in someembodiments, the differential switched signal 86 a, 86 b can cancel someother undesirable signal components within the primary circuit path 112.

In some embodiments, the integrators 78, 84 are continuous, i.e.,un-sampled, integrators, which can be either active or passive. Bothactive and passive integrator structures are known.

In other embodiments, the integrators 78, 84 can be switched capacitorintegrators described more fully below in conjunction with FIG. 5B.Switched capacitor integrators require clock signals. Thus, clocksignals 98, 99, 102, 103 are shown to be received by the integrators 78,84, respectively. The clock signals 98, 99, 102, 103 can be at anyfrequency, fx. However, as is known, any sampled system generates nullsin their transfer function at a clock frequency and at higher harmonicsthereof. Therefore, it may be desirable to select a frequency of theclock signals 98, 99, 102, 103 such that nulls occur at particularfrequencies, for example, at frequencies of fchop or fchop/2. The clocksignals 98, 99, 102, 103 are described more fully below in conjunctionwith FIG. 5C.

The switching circuits 82, 86 can be controlled by clock signals 100,104, respectively, with frequencies of 2fchop/N. The switching circuit76 can be controlled by a clock signal 96 with a frequency of fchop.Reasons for the selection of particular clock frequencies will becomemore apparent below in conjunction with FIGS. 6-23.

Operation of the magnetic field sensor 50 is described in conjunctionwith FIGS. 6-23 below.

Referring now to FIG. 4, in which like elements of FIG. 3 are shownhaving like reference designations, another magnetic field sensor 120can include the chopping circuit 110, the primary circuit path 112, andthe gain feedback circuit 114. However, here shown, the differentialoutput signal 66 a′, 66 b′ is shown with prime symbols to indicate thatthe differential output signal 66 a′, 66 b′ is very much like thedifferential output signal 66 a, 66 b of FIG. 3, but also that itdiffers slightly due to differences in feedback circuits describedbelow. For similar reasons, the gain feedback signal 70 a′, 70 b′ isshown with prime symbols.

The magnetic field sensor 120 can include a first feedback circuit 126and an Mth feedback circuit 127, which together form of feedback circuitpath coupled at both ends to the primary circuit path 112 to form afeedback loop. As described above in conjunction with FIG. 3, M=N/2.

The first feedback circuit 126 is similar to the feedback circuit 116 ofFIG. 3. However, the first feedback circuit 126 does not includecapacitors 72, 74, and it also includes a different type of integrator.

The first feedback circuit 126 includes a switching circuit 122 coupledto receive the differential output signal 66 a′, 66 b′ and configured togenerate a differential switched signal 122 a, 122 b, which is labeledas a signal N. An N-phase switched capacitor notch filter integrator 124is coupled to receive the differential switched signal 122 a, 122 b andconfigured to generate a differential integrated signal 124 a, 124 b,which is labeled as a signal R. Examples of switched capacitor notchfilter integrators can be found, for example, in U.S. Pat. No.7,990,209, issued Aug. 2, 2011, assigned to the assignee of the presentinvention, and incorporated by reference herein in its entirety. Also,an example of a switched capacitor notch filter integrator is shown anddescribed below in conjunction with FIGS. 5, 5A, and 5D.

In general, notches in the transfer function of a switched capacitornotch filter integrator can be controlled by a separate clock signal,i.e., a redistribution or averaging clock signal, apart from a primarysampling clock signal. Thus, the redistribution clock provides anenhanced ability to position notches in a corresponding transferfunction, as further described below.

The summing circuit 58 can be coupled to receive the signal 124 b andthe summing circuit 60 can be coupled to receive the signal 124 a, orvice versa.

The Mth feedback circuit 127 can include a switching circuit 128 coupledto receive the differential switched signal 122 a, 122 b and configuredto generate a differential switched signal 128 a, 128 b, which islabeled as a signal O.

An N-phase switched capacitor notch filter integrator 130 can be coupledto receive the differential switched signal 128 a, 128 b and configuredto generate a differential integrated signal 130 a, 130 b, which islabeled as a signal P. A switching circuit 132 can be coupled to receivethe differential integrated signal 130 a, 130 b and configured togenerate a differential switched signal 132 a, 132 b, which is labeledas a signal Q.

The summing circuit 158 can be coupled to receive the signal 132 b andthe summing circuit 60 can be coupled to receive the signal 132 a.

The switched capacitor notch filter integrators 124, 130 can be coupledto receive sample clock signals 134, 138, respectively at a frequency of2fchop/N and also redistribution clock signals 136, 140, respectively,at a frequency of 2fchop/N. The sample clock signals 134, 140 are eachcomprised of four clock signals at different phases. The four differentphases are described more fully below in conjunction with FIGS. 5 and5A.

The switching circuits 128, 132 can be switched with clock signals 138,144, respectively, at a frequency of 2fchop/N. The switching circuit 122can be switched with a clock signal 133 at a frequency of fchop. Reasonsfor the selection of frequencies will become more apparent below inconjunction with FIGS. 6-23.

Operation of the magnetic field sensor 120 is described in conjunctionwith FIGS. 6-23 below.

The N-phase switched capacitor notch filter integrators 124, 130 of FIG.4 have particular advantages over simple integrators 78, 84 (linear orswitched capacitor types) described above in conjunction with FIG. 3. Itwill be appreciated from discussion below in conjunction with spectralplots in FIGS. 6-23, that an ability to position notches of the N-phaseswitched capacitor notch filter integrators 124, 130 (e.g., by way ofthe redistribution clock signals described above) provides the abilityto remove signal components that are not intended to be integrated.Accordingly, it will be appreciated that the AC coupling capacitors 72,74 of FIG. 3 are not required. Furthermore, the ability to place notchesat frequencies selected to remove all spectral lines that are notintended to be integrated (i.e., undesirable spectral components) avoidsusing an integrator with a very low cut-off integration frequency, whichwould otherwise be required in order to achieve a very large attenuationof those undesirable spectral components. Therefore, the equivalence oflarge capacitors (for achieving very low cut-off frequencies) can beachieved, but without large capacitors. Having higher cut-offfrequencies allows for the feedback circuits 126, 127 to settle morerapidly, resulting in a rapid removal of the undesirable spectralcomponents.

Referring now to FIG. 5, a four-phase switched capacitor notch filterintegrator 150 is representative of the N-phase switch capacitor notchfilter integrator 130 of FIG. 4 for the case of 4× chopping, where N=4.The switched capacitor notch filter integrator 150 is coupled to receivea differential input signal 152 a, 152 b. The switched capacitor notchfilter integrator 150 includes a plurality of switches and a pluralityof capacitors 154 all coupled is shown, and coupled to input nodes of anamplifier 156. Some of the plurality of switches are controlled by aso-called “redistribution” clock signal, CPR. Other ones of theplurality of switches are controlled by a sample clock signals, CP1,CP2, CP3, CP4. Each one of the sample clock signals has the samefrequency but occurs a different phase as described more fully below inconjunction with FIG. 5A.

The amplifier 156 is configured to generate a differential output signal156 a, 156 b, which can be the same as or similar to the differentialoutput signal 130 a, 130 b of FIG. 4.

A capacitor 158 is coupled between an input node of the amplifier 156and the output signal 156 a. A capacitor 160 is coupled between anotherinput node of the amplifier 156 and the output signal 156 b.

Referring now to FIG. 5A, a graph 170 has a horizontal axis with a scalein units of time in arbitrary units and a vertical axis with a scale inunits of volts in arbitrary units. A signal 172 can be the same as orsimilar to the differential input signal 152 a, 152 b of FIG. 5.

Clock signals 174, 176, 178, 180 can be the same as or similar to thesampling clock signals, CP1, CP2, CP3, CP4 of FIG. 5 and the clocksignals 140 of FIG. 4. The clock signals 174, 176, 178, 180 are shown aspluralities of dark boxes representing sampling periods, however, thedark boxes are representative of switch closures of respective ones ofthe pairs of switches in FIG. 5. As described above the sample clocksignal 140 is actually four clock signals, each at the same frequency,and each at a different phase.

A clock signal 182 can be the same as or similar to the redistributionor averaging clock signals 136, 142 of FIG. 4. The clock signal 182 isshown as a plurality of dark boxes, however, the dark boxes arerepresentative of switch closures of respective ones of the switches inFIG. 5. In operation, at times when the redistribution clock signal 182is high, samples associated with each one of the sample clock signals174, 176, 178, 180 are averaged.

Referring now to FIG. 5B, a conventional switched capacitor integratorcan be the same as or similar to the integrators 78, 84 of FIG. 3. Theswitched capacitor integrator 200 is coupled to receive a differentialinput signal 202 a, 202 b. The switched capacitor integrator 200includes a plurality of switches and a plurality of capacitors 204 allcoupled is shown, and coupled to input nodes of an amplifier 206. Someof the plurality of switches are controlled by a redistribution clocksignal, CKR. Other ones of the plurality of switches are controlled by asample clock signal, CS. In some embodiments, the sample clock signaland the redistribution clock signal have the same frequency butdifferent phases.

The amplifier 206 is configured to generate a differential signal 206 a,206 b, which can be the same as or similar to the differential signals78 a, 78 b and 84 a, 84 b of FIG. 3.

A capacitor 208 is coupled between an input node of the amplifier 206and the output signal 206 a. A capacitor 210 is coupled between anotherinput node of the amplifier 206 and the output signal 206 b.

Referring now to FIG. 5C, a graph 220 has a horizontal axis with a scalein units of time in arbitrary units and a vertical axis with a scale inunits of volts in arbitrary units. A signal 222 can be the same as orsimilar to the differential input signal 202 a, 202 b of FIG. 5B.

A clock signal 224 can be the same as or similar to the clock signal,CS, of FIG. 5B and the clock signals 98, 102 of FIG. 3. The clocksignals 224 is shown as dark boxes, however, the dark boxes arerepresentative of switch closures of respective ones of the switches inFIG. 5B.

A clock signal 226 can be the same as or similar to the redistributionor averaging clock signals 99, 103 of FIG. 3. The clock signal 226 isshown as dark boxes, however, the dark boxes are representative ofswitch closures of respective ones of the switches in FIG. 5B.

Referring now to FIG. 5D, a graph 240 includes a horizontal axis inunits of frequency and a vertical axis in non-dimensional units. A curve242 is indicative of a transfer function, and, in particular, a transferfunction of the four-phase notch filter switched capacitor integrator150 of FIG. 5 and, in four phase chopping arrangements, of the N-phasenotch filter switched capacitor integrator 130 of FIG. 4. It will beunderstood that the transfer function of this integrator, and of anyintegrator, has a high gain at DC, and rolls off at higher frequencies.This particular transfer function rolls off at frequencies above DC at arate generally of about 20 dB per decade. The transfer function 242 isrepresentative of a sin x/x (or sinc) function. A first null is shown ata frequency of fchop/2.

It will be understood that a frequency of the first null can becontrolled by the redistribution clock signal 182 of FIG. 5A, theredistribution clock signal, CKR, of FIG. 5, and the redistributionclock signal 142 of FIG. 4.

It will also be understood that the curve 242 has a shape generallyrepresentative of a closed loop transfer function, for example, a closedloop transfer function of the feedback circuits 126, 127 of FIG. 4.However, the closed loop transfer function will generally have a lowergain at DC than an open loop transfer function of the N-phase notchfilter switched capacitor integrator 130.

Similar circuits and transfers functions can be used for any of theN-phase notch filter switched capacitor integrators of FIG. 4.

FIGS. 6-23 show frequency domain graphs representative of signalslabeled A-R in FIGS. 3 and 4, for the case of 4× chopping of the HallEffect element 52.

In each graph, three spectral lines are shown at three respectivedifferent frequencies and amplitudes. For clarity, the spectral linesare representative of the magnetic field sensors of FIGS. 3 and 4 beingexposed to an extra magnetic field with only a DC magnetic fieldcomponent. However, if the magnetic field sensors are exposed to an ACexternal magnetic field, each one of the spectral lines will broadeninto spectral bands.

At various points in the circuit three of FIGS. 3 and 4, and in variouscorresponding ones of FIGS. 6-23, positions of the three spectral lineschange. However, the spectral lines can be identified by way of theirrespective amplitudes no matter at what frequency or position they arefound. Gain of amplifiers is not represented in FIGS. 6-23 for clarity,in order to keep the spectral lines at the same amplitude so that theycan be readily identified.

In some of the graphs, spectral lines are shown as dashed lines ratherthan solid lines. The dashed lines indicate that those spectral lineschange with time for a time period beginning at a power up of themagnetic field sensors 50, 120 of FIGS. 3 and 4, respectively. Some ofthe dashed spectral lines occur at full magnitude when the magneticfield sensors 50 and 120 of FIGS. 3 and 4 first power up, andthereafter, the dashed spectral lines diminish toward zero amplitude.Others of the dashed spectral lines occur at very low magnitude when themagnetic field sensors 50 and 120 of FIGS. 3 and 4 first power up, andthereafter, the dashed spectral lines increase in magnitude. This effectis described more fully below in conjunction with FIGS. 24-26.

Referring now to FIG. 6, the signal labeled A in FIGS. 3 and 4, due tochopping of the Hall element 52, has three spectral lines (an alsohigher order spectral lines, not shown, but at lower amplitudes). Aspectral line at a frequency, fc=fchop, is a desired signal,Bext+Resoff. The desired signal is representative of an externalmagnetic field, here a DC magnetic field, sensed by the magnetic fieldsensors of FIGS. 3 and 4.

Spectral lines within the signal labeled A at DC and at a frequency,fc/2, are undesirable. The spectral line at DC is representative of oneaspect of Hall element DC offset, HP Off2X (see, e.g., FIG. 2, signal20). The spectral line at a frequency fc/2 is representative of anotheraspect of Hall element DC offset, HP Off4X (see, e.g., FIG. 2, signal24).

Referring now to FIG. 7, the signal labeled B in FIGS. 3 and 4 hasessentially the same spectral content as the signal labeled A. Thesignal labeled A has passed through the amplifier 56 of FIGS. 3 and 4 toresult in the signal labeled B. As described above, gain of amplifiersis not included in the graphs.

Referring now to FIG. 8, the signal labeled C in FIGS. 3 and 4 also hasthe same spectral content as the signals labeled A and B. However, thesignal labeled C experiences spectral lines that change in amplitude fora time period after power up, i.e., spectral lines at DC and at afrequency of fc/2.

Referring now to FIG. 9, the signal labeled D in FIGS. 3 and 4 also hasthe same spectral content as the signals labeled A, B, and C. The signallabeled C has passed though one more amplifier 62 to result in thesignal labeled D. As described above, gain of amplifiers is not includedin the graphs.

Referring now to FIG. 10, the signal labeled E in FIGS. 3 and 4 hasdifferent spectral content than the signal labeled D. The signal labeledD has passed through the switching circuit 64 to result in the signallabeled E. The switching circuit 64 operates to multiplex the signal Ewith the clock signal 92 at a frequency of fc=fchop. As is known, themultiplexer generates sum and difference products, resulting in thespectral content shown, in which positions of spectral lines havechanged.

Referring now to FIG. 11, the signal labeled F in FIGS. 3 and 4 has thesame spectral content as the signal labeled E. The signal labeled E haspassed though one more amplifier 66 to result in the signal labeled F.As described above, gain of amplifiers is not included in the graphs.

Referring now to FIG. 12, the signal labeled G in FIG. 3 has differentspectral content than the signal labeled F. The signal labeled F haspassed through capacitors 72, 74 to result in the signal labeled G. Thecapacitors remove the DC component. In some alternate embodiments, thecapacitors 72, 74 are not used.

Referring now to FIG. 13, the signal labeled H in FIG. 3 has differentspectral content than the signal labeled G. The signal labeled G haspassed through the switching circuit 76 to result in the signal labeledH. The switching circuit 76 operates to multiplex the signal labeled Gwith the clock signal 96 at a frequency of fchop, resulting in thespectral content shown, in which positions of spectral lines havechanged.

Referring now to FIG. 14, the signal labeled I in FIG. 3 has differentspectral content than the signal labeled H. The signal labeled H haspassed through the switching circuit 82 to result in the signal labeledI. The switching circuit 82 operates to multiplex the signal labeled Hwith the clock signal 100 at a frequency of 2fc/N, which is fc/2 for thecase of 4× chopping, resulting in the spectral content shown, in whichpositions of spectral lines have changed.

Referring now to FIG. 15, the signal labeled J in FIG. 3 has differentspectral content than the signal labeled I. The signal labeled I haspassed through the integrator 84, which has high gain at DC and low gainat other frequencies, to result in the signal labeled J. Thus, primarilythe DC component of the signal labeled I remains in the signal labeledJ.

Referring now to FIG. 16, the signal labeled K in FIG. 3 has differentspectral content than the signal labeled J. The signal labeled J haspassed through the switching circuit 86 to result in the signal labeledK. The switching circuit 86 operates to multiplex the signal labeled Jwith the clock signal 100 at a frequency of 2fc/N, which is fc/2 for thecase of 4× chopping, resulting in the spectral content shown, in whichpositions of spectral lines have changed, and only one spectral lineremains, at a frequency of fc/2. The signal labeled K is summed back(see summation circuits 58, 60 of FIG. 3) into the signal labeled B ofFIG. 3 at the proper phase to reduce or eliminate the spectral line inthe signal labeled C at the frequency of fc/2.

Referring now to FIG. 17, the signal labeled L in FIG. 3 has differentspectral content than the signal labeled H by operation of theintegrator 78, for reasons described above. The signal labeled L hasspectral content primarily at DC. The signal labeled L is summed back(see summation circuits 58, 60 of FIG. 3) into the signal labeled B ofFIG. 3 at the proper phase to reduce or eliminate the spectral line inthe signal labeled C at DC.

By way of the signals labeled K and L, both of the undesirable spectrallines in the signal labeled C of FIG. 3 are progressively removed duringa time period following power up, leaving the desired spectral line atthe frequency of fc in the signal labeled C. It should be appreciatedthat the signal labeled F in FIG. 3 can be the output signal from themagnetic field sensor 50 of FIG. 3. Note that in FIG. F, the position ofthe desired spectral line has changed to DC.

As described above, the graphs of FIGS. 6-23 are representative of themagnetic field sensors of FIGS. 3 and 4 being exposed to a DC externalmagnetic field. If instead, the magnetic field sensors were exposed toan AC magnetic field, all of the spectral lines shown in the graphswould broaden the spectral bands. Thus, for the case of an AC externalmagnetic field, the remaining spectral line, shown to be at DC, wouldinstead be a spectral band centered at DC.

Referring now to FIG. 18, the signal labeled M in FIGS. 3 and 4 hasdifferent spectral content than the signal labeled F. The signal labeledF has passed through the switching circuit 70 to result in the signallabeled M. The switching circuit 70 operates to multiplex the signallabeled F with the clock signal 94 at a frequency of fc, resulting inthe spectral content shown, in which positions of spectral lines havechanged.

FIGS. 19-23 are representative of signals that appear only in FIG. 4.

Referring now to FIG. 19, the signal labeled N in FIG. 4 has differentspectral content than the signal labeled F. The signal labeled F haspassed through the switching circuit 122 to result in the signal labeledN. The switching circuit 122 operates to multiplex the signal labeled Fwith the clock signal 133 at a frequency of fc, resulting in thespectral content shown, in which positions of spectral lines havechanged

Referring now to FIG. 20, the signal labeled O in FIG. 4 has differentspectral content than the signal labeled N. The signal labeled N haspassed through the switching circuit 128 to result in the signal labeledO. The switching circuit 128 operates to multiplex the signal labeled Nwith the clock signal 138 at a frequency of 2fc/N, which is fc/2 for thecase of 4× chopping, resulting in the spectral content shown, in whichpositions of spectral lines have changed.

Referring now to FIG. 21, the signal labeled P in FIG. 4 has differentspectral content than the signal labeled O. The signal labeled O haspassed through the N-phase (here four-phase) switched capacitor notchfilter integrator 130, which has high gain at DC and low gain at otherfrequencies, to result in the signal labeled P. Thus, primarily the DCcomponent of the signal labeled O remains in the signal labeled P.

Referring now to FIG. 22, the signal labeled Q in FIG. 4 has differentspectral content than the signal labeled P. The signal labeled P haspassed through the switching circuit 132 to result in the signal labeledQ. The switching circuit 132 operates to multiplex the signal labeled Pwith the clock signal 144 at a frequency of 2fcN, which is fc/2 for thecase of 4× chopping, resulting in the spectral content shown, in whichpositions of spectral lines have changed.

Referring now to FIG. 23, the signal labeled R in FIG. 4 has differentspectral content than the signal labeled N. The signal labeled R haspassed through the N-phase (here two-phase) switched capacitor notchfilter integrator 124, which has high gain at DC and low gain at otherfrequencies. Thus, primarily the DC component of the signal labeled Nremains in the signal labeled R.

By way of the signals labeled Q and R, both of the undesirable spectrallines in the signal labeled C of FIG. 4 are progressively removed,leaving the desired spectral line at the frequency of fc. It should beappreciated that the signal labeled F of FIG. 4 can be the output signalfrom the magnetic field sensor 120 of FIG. 4. Note that in the signallabeled F, the position of the desired spectral line has moved to DC.

FIGS. 24-26 show the above-described start up (i.e., power up) behaviorof the magnetic fields sensors 50, 120 of FIGS. 3 and 4, respectively,for the case where the magnetic field sensors are exposed to a DCmagnetic field. FIGS. 23-24 of each show graphs having horizontal axeswith scales in units of time and vertical axes with scales in units ofvoltage.

Referring now to FIG. 24, a signal is representative of the signallabeled F of FIGS. 3 and 4, i.e., the output signal of the magneticfield sensors. It can be seen that the output signal has undesirablespectral content for some time period beginning at time zero (i.e.,power up) and extending out to about 0.4 milliseconds. This time periodis determined by a variety factors, for example, transfer functioncharacteristics of the integrators of FIGS. 3 and 4. It is desirablethat this time be short.

Referring now to FIG. 25, a signal is representative of the signalslabeled L and R of FIGS. 3 and 4, respectively. As described above, thesignals labeled L and R are DC feedback signals, and they increase withtime so as to cancel undesirable DC spectral components in the primarycircuit path 112 of FIGS. 3 and 4.

Referring now to FIG. 26, a signal is representative of the signalslabeled K and Q of FIGS. 3 and 4, respectively. As described above, thesignals labeled K and Q have spectral content at a frequency of fc/2,and they increase with time so as to cancel undesirable spectralcomponents at the frequency of fc/2 in the primary circuit path 112 ofFIGS. 3 and 4.

It will be appreciated that, by using techniques described above toremove undesirable spectral components, the primary circuit path 112 ofFIGS. 3 and 4 does not need to use filters. As described above, filterstend to slow the response time of magnetic field sensors. Thus, usingthe circuits and techniques described above, magnetic field sensors canbe built that have faster response times.

All references cited herein are hereby incorporated herein by referencein their entirety.

Having described preferred embodiments, which serve to illustratevarious concepts, structures and techniques, which are the subject ofthis patent, it will now become apparent to those of ordinary skill inthe art that other embodiments incorporating these concepts, structuresand techniques may be used. Accordingly, it is submitted that that scopeof the patent should not be limited to the described embodiments butrather should be limited only by the spirit and scope of the followingclaims.

What is claimed is:
 1. A magnetic field sensor, comprising: a magnetic field sensing element configured to generate an electronic signal in response to a magnetic field; an N-phase modulator coupled to receive the electronic signal and configured to generate a modulated signal having a plurality of frequency components at different frequencies, wherein at least two one of the plurality of frequency components correspond to at least two undesirable frequency components and one of the plurality of frequency components corresponds to a desirable frequency component, wherein the desirable frequency component comprises a magnetic field signal representative of the magnetic field; a primary circuit path coupled to receive the modulated signal and to process the modulated signal to generate an output signal representative of the magnetic field signal; and a feedback circuit path coupled at both ends of the feedback circuit path to the primary circuit path and forming a feedback loop, the feedback circuit path configured to generate a feedback signal and coupled to add the feedback signal to the primary circuit path to cancel the at least two undesirable frequency components.
 2. The magnetic field sensor of claim 1, wherein, during a first time period, the output signal from the primary circuit path is representative of both the desirable frequency component and of the at least two undesirable frequency components, and during a second different time period after the first time period, the output signal from the primary circuit path is primarily representative of the desirable frequency component.
 3. The magnetic field sensor of claim 1, wherein the feedback circuit path comprises: a first one or more modulators coupled in series and coupled to receive the output signal from the primary circuit path and configured to generate a modulated signal having one undesirable frequency component of the at least two undesirable frequency components at baseband; an integrator coupled to receive the modulated signal and configured to generate an integrated signal related to the feedback signal.
 4. The magnetic field sensor of claim 3, wherein the integrator comprises a switched capacitor integrator.
 5. The magnetic field sensor of claim 3, wherein the integrator comprises an N-phase switched capacitor notch filter integrator.
 6. The magnetic field sensor of claim 3, wherein the integrator comprises a continuous linear integrator.
 7. The magnetic field sensor of claim 3, wherein the feedback circuit path further comprises a second one or more modulators coupled in series and coupled to receive the integrated signal and configured to shift a frequency of the integrated signal to generate a frequency shifted signal related to the feedback signal.
 8. The magnetic field sensor of claim 7, wherein the magnetic field sensing element comprises a Hall effect element, and wherein the N-phase modulator comprises a chopping circuit.
 9. The magnetic field sensor of claim 8, wherein the chopping circuit comprises a 4× chopping circuit.
 10. The magnetic field sensor of claim 1, wherein the plurality of frequency components comprises a plurality of undesirable frequency components and also the desirable frequency component, wherein the feedback circuit path comprises a plurality of feedback circuits, each feedback circuit coupled at both ends to the primary circuit path and forming a respective feedback loop, each feedback circuit configured to generate a respective feedback signal, each feedback circuit coupled to add the respective feedback signal to the primary circuit path to cancel a different respective one of the plurality of undesirable frequency components.
 11. The magnetic field sensor of claim 10, wherein, during a first time period, the output signal from the primary circuit path is representative of both the desirable frequency component and of the plurality of undesirable frequency components, and during a second different time period after the first time period, the output signal from the primary circuit path is primarily representative of the desirable frequency component.
 12. The magnetic field sensor of claim 10, wherein each one of the plurality of feedback circuits comprises: a respective first one or more modulators coupled in series and coupled to receive the output signal from the primary circuit path and configured to generate a respective modulated signal having a respective one of the plurality of undesirable frequency components at baseband; and a respective integrator coupled to receive the respective modulated signal and configured to generate a respective integrated signal related to the respective feedback signal.
 13. The magnetic field sensor of claim 12, wherein each respective integrator comprises a respective switched capacitor integrator.
 14. The magnetic field sensor of claim 12, wherein each respective integrator comprises a respective N-phase switched capacitor notch filter integrator.
 15. The magnetic field sensor of claim 12, wherein each respective integrator comprises a respective continuous linear integrator.
 16. The magnetic field sensor of claim 12, wherein each one of the plurality of feedback circuit paths further comprises a respective second one or more modulators coupled in series and coupled to receive the respective integrated signal and configured to shift a frequency of the respective integrated signal to generate a respective frequency shifted signal related to the respective feedback signal.
 17. The magnetic field sensor of claim 16, wherein the magnetic field sensing element comprises a Hall effect element, and wherein the N-phase modulator comprises a chopping circuit.
 18. The magnetic field sensor of claim 17, wherein the chopping circuit comprises a 4× chopping circuit.
 19. A method of processing a signal in a magnetic field sensor, comprising: generating a magnetic field signal with a magnetic field sensor in response to a magnetic field; modulating the magnetic field signal with an N-phase modulator coupled to receive the magnetic field signal and configured to generate a modulated signal having a plurality of frequency components at different frequencies, wherein at least two of the plurality of frequency components correspond to at least two undesirable frequency components and one of the plurality of frequency components corresponds to a desirable frequency component; processing the modulating signal with a primary circuit path coupled to receive the modulated signal, the primary circuit path configured to generate an output signal representative of the desirable frequency component; generating a feedback signal with a feedback circuit path; and adding the feedback signal to the primary circuit path to cancel the at least two undesirable frequency components.
 20. The method of claim 19, wherein, during a first time period, the output signal from the primary circuit path is representative of both the desirable frequency component and of the at least two undesirable frequency components, and during a second different time period after the first time period, the output signal from the primary circuit path is primarily representative of the desirable frequency component.
 21. The method of claim 19, wherein the generating the feedback signal comprises: generating, with a first one or more modulators coupled in series, a modulated signal having one undesirable frequency component of the at least two undesirable frequency components at baseband; generating, with an integrator, an integrated signal related to the feedback signal.
 22. The method of claim 21, wherein the integrator comprises a switched capacitor integrator.
 23. The method of claim 21, wherein the integrator comprises an N-phase switched capacitor notch filter integrator.
 24. The method of claim 21, wherein the integrator comprises a continuous linear integrator.
 25. The method of claim 21; further comprising: shifting, with a second one or more modulators, a frequency of the integrated signal to generate a frequency shifted signal related to the feedback signal.
 26. The method of claim 19, wherein the plurality of frequency components comprises a plurality of undesirable frequency components and also the desirable frequency component, wherein the feedback circuit path comprises a plurality of feedback circuits, and wherein the generating the feedback signal comprises: generating a respective feedback signal with each one of the plurality of feedback circuits; and adding each respective feedback signal to the primary circuit path to cancel a different respective one of the plurality of undesirable frequency components.
 27. The method of claim 26, wherein, during a first time period, the output signal from the primary circuit path is representative of both the desirable frequency component and of the plurality of undesirable frequency components, and during a second different time period after the first time period, the output signal from the primary circuit path is primarily representative of the desirable frequency component.
 28. The method of claim 26, wherein each one of the plurality of feedback circuits comprises: a respective first one or more modulators coupled in series and coupled to receive the output signal from the primary circuit path and configured to generate a respective modulated signal having a respective one of the plurality of undesirable frequency components at baseband; and a respective integrator coupled to receive the respective modulated signal and configured to generate a respective integrated signal related to the respective feedback signal.
 29. The method of claim 28, wherein each respective integrator comprises a respective switched capacitor integrator.
 30. The method of claim 28, wherein each respective integrator comprises a respective N-phase switched capacitor notch filter integrator.
 31. The method of claim 28, wherein each respective integrator comprises a respective continuous linear integrator.
 32. The method of claim 28, wherein each one of the plurality of feedback circuit paths further comprises a respective second one or more modulators coupled in series and coupled to receive the respective integrated signal and configured to shift a frequency of the respective integrated signal to generate a respective frequency shifted signal related to the respective feedback signal. 